1. Introduction

Along with the fast growth in the display market with recent digital multimedia broadcasting (DMB) age, power engineers have been studying on efficient power driving methods for the backlight unit (BLU) of LCD TV. In respect of power electronics, especially, compact and cost-effective topologies and the accompanying EMI reduction are strongly required for low-power supplies of less than 100 W for the low and middle-priced TV market [1, 2]. A flyback converter has been generally applied to the switched-mode power supply (SMPS) for 75-W TVs. However, the flyback converter needs a snubber circuit to suppress high voltage stress across the switch caused by the leakage inductance of the transformer. Besides, high di/dt and dv/dt by the hard switching result in increasing costs to solve EMI problems, and increased switching losses lead to reduced overall efficiency and limited operation of the converters at high frequency. In order to avoid these drawbacks, soft-switching techniques can be considered as an effective solution. These techniques are classified into two types of topologies: constant switching frequency and variable frequency converters [3-15]. However, the converters with constant frequency employ extra components for the soft switching that cause complex control and cost increases [3-8]. Some variable frequency converters have been proposed. The LLC- or SRC-typed resonant converter enables switches to use ZVS operation and promote system efficiency [9-13]. However, these converters employ switches more than two for bridge configurations, which is more appropriate for TV applications at more than 150 W. Alternatively, the quasi-resonant converters with only a single switch can be valid topologies [14, 15]. However, the main switch in one approach does not guarantee the soft switching, and the free-wheeling diode in the secondary side is hard switched at turn-on [14]. Moreover, the variation of the switching frequency is more than about 200 kHz for load changes. On the other hand, in another approach [15], the main switch and the rectifying diode are stressed by the resonant current during the turn-on time.

In this paper, a single-switch ZVZCS quasi-resonant CLL isolated DC-DC converter applicable for low-power applications is proposed. The operation at the switch turn-on is identical to the forward converter. Even though the switch is turned off, the stored energy of the resonant inductor in the primary side is transferred to the load during a certain period. Therefore, this converter has the operational characteristics of both forward and flyback converters without the output filter inductor. In addition, all of semiconductors in the converter are soft switched at turn-on and turn-off as well as the switching frequency of the proposed converter slightly varies for loads unlike the conventional variable frequency converters. Consequently, the efficiency reduction under light load is minimized. For conducted emissions higher than 1 MHz, the proposed converter shows considerable advantages compared with the conventional flyback converter.

In Section II, four operating modes about the proposed converter are explained, and the detailed characteristic analysis and control strategy are described. To verify the theoretical analysis, the experimental results are presented in Section III. The conclusion is provided in Section IV.

2. Proposed Converter

2.1 Description of the Operation Modes

Fig. 1 shows the circuit model of the proposed converter. It consists of a single-switch SW, a resonant capacitor Cr and inductor Lr, a rectifying diode D, and a transformer, which has a magnetizing inductance Lm and a turn ratio of n : 1. To simplify the analysis, it is assumed that:

All components are ideal and the converter is operating in steady state. The input voltage Vin and the output voltage Vo are constant during the switching period because the capacitor is sufficiently large. The leakage inductance of the transformer is neglected.

Fig. 1.Circuit model of the proposed converter

The steady-state operation includes four modes in one switching period TS. The operating modes and waveforms are shown in Figs. 2 and 3, respectively.

Fig. 2.Operating modes of the proposed converter

Fig. 3.Operating waveforms of the proposed converter

1) Mode 1 [t0, t1]: At t4, when the capacitor voltage vCr reaches the input voltage Vin and the switch voltage vSW becomes zero, the anti-parallel diode of the switch begins to conduct. During this interval, the switch can achieve the ZVS, so that its turn-on switching loss is eliminated. Also, the rectifying diode can turn on under zero-current-switching (ZCS) conditions, because its current id increases from zero.

2) Mode 2 [t1, t2]: Prior to t1, the main switch SW is turned on. The current through the resonant inductor Lr and the magnetizing inductance Lm linearly increase from zero until t2, because the voltages across the resonant capacitor Cr and the magnetizing inductance Lm are Vin and nVo, respectively. The converter transfers the input power to the load through the transformer like a forward converter. The switch current iSW is the same as the resonant inductor current iLr. During this mode, the resonant inductor current iLr and the magnetizing current iLm are represented as follows:

where nVo is the output voltage reflected to the primary side of the transformer.

3) Mode 3 [t2, t3]: After the switch SW is turned off at t2, the inductor Lr commences resonance with the resonant capacitor Cr. The switch voltage vSW increases smoothly by the resonance, leading it to be turned off under ZVS. Even though the switch was turned off, the stored energy in the resonant inductor Lr is transferred to the load like a flyback converter. The voltage across the magnetizing inductance Lm is clamped to nVo so that the magnetizing current iLm still flows linearly, as shown in Fig. 3. During this mode, the resonant current iLr, the magnetizing current iLm, and the resonant capacitor voltage vCr are obtained as follows:

where the characteristic impedance Zo1 and the angular frequency ωo2 in this mode are given by:

4) Mode 4 [t3, t4]: At t3, the secondary rectifying diode D finishes conducting under almost ZCS conditions when the resonant current iLr coincides with the magnetizing current iLm. In this mode, the resonant capacitor Cr is charged and discharged through Lr and Lm until its voltage reaches the input voltage Vin at t4. The resonant inductor current iLr is the same as the magnetizing inductor current iLm. The resonant current iLr and the resonant capacitor voltage vCr are shown as follows:

where the characteristic impedance Zo1 and the angular frequency ωo2 in this mode are given by:

Although the voltage stresses across the switch and the rectifying diode increase by the resonance between the resonant inductor Lr, the resonant capacitor Cr, and the magnetizing inductor Lm during mode 4, the above description of the proposed converter operation shows that the main switch SW and the rectifying diode D are always turned on and turned off under soft-switching conditions, thereby eliminating the switching losses.

2.2 Analysis of the operation characteristics

In this section, the operational characteristics of the proposed converter are discussed. Firstly, the input to output voltage gain of the converter is defined, and variables related to the voltage gain are analyzed.

This converter operates as a forward converter during switch turn-on. On the other hand, after switch turn-off, the energy stored in the resonant inductor Lr and capacitor Cr is transferred to the load like a flyback converter until the resonant inductor current iLr is the same as the magnetizing inductor current iLm. Fig. 4 shows current waveforms of each component during the rectifying diode conduction for an arbitrary load. The magnetizing inductor current iLm subtracted from the resonant inductor current iLr is multiplied by the transformer turns ratio n. It consists of the rectifying diode current. The average value of the periodic signal can be solved by dividing the total area by the period. In order to calculate the average current of the rectifying diode, the variable K is introduced. K is defined as the ratio of conduction areas during tON and t23 for the given circuit parameters and is given by:

where tON is the conduction time during modes 1 and 2, and t23 is the conduction time during mode 3.

Fig. 4.Key current waveforms during the diode conduction

In steady-state operation, when the input power begins to transfer to the load after mode 4, the initial values of the resonant inductor current iLr and magnetizing inductor current iLm are always identical, and its currents increase linearly. Therefore, by using (1) and (2), the rectifying diode peak current IDpk during switch turn-on can be expressed by:

From (10) and (11), AON and the average current of the rectifying diode ID are derived as follows:

Eventually, because the average current of the rectifying diode is equal to the output current, the input to output voltage gain of the proposed converter can be characterized as functions of the quality factor Q, normalized frequency fn, inductance ratio λ, transformer turns ratio n, and duty ratio D(= tON /TS):

The proposed converter is able to operate in no-load conditions. When Q = 0, the voltage gain of the proposed converter is expressed as a function of the transformer turn ratio n and the inductance ratio λ:

Then, A23 should be calculated. In (10), AON can be more easily calculated using (11) and (12). On the other hand, in order to calculate A23, it is necessary to know the initial values of iLr and iLm at the beginning of mode 3. In modes 1 and 2, the voltage across Lm is clamped to nVo. Therefore, the absolute value of iLm at turn-on of the anti-parallel diode of the switch is the same as the value of iLm at switch turn-off. Also, because the switch current iSW flows linearly through the resonant inductor Lr and the magnetizing inductor Lm, the initial value of the resonant inductor current iLr coincides with the initial value of the magnetizing inductor current iLm at t = t0. Each initial value of (1), (2), and (4) is defined by:

At t = t2, the resonant inductor current iLr can be derived as (18) by substituting (17) into (1):

Therefore, because the energy stored in the resonant capacitor Cr and inductor Lr during tON is transferred to the load in mode 3, A23 can be derived by integrating the rectifying diode current iD on t23 by using (3), (4), (17), and (18):

A23 is a relatively small portion of the total area compared to AON. In order to facilitate the analysis, a triangle area is considered for estimating A23. Eq. (19) can be simplified as:

From (12) and (20), the area ratio factor K described earlier is rearranged as

The average current of the rectifying diode ID is approximately determined by the values of the resonant inductor current iLr and the magnetizing inductor current iLm at t = tON and the resonant time t23 between Cr and Lr. In the steady-state operation, the currents iLr and iLm are always the same at the end of mode 3. Therefore, the time t23 can be calculated through (3), (4), (17), and (18):

In (22), the time t23 is constant regardless of loads for the given resonant parameters when the input to output voltage gain is decided.

For a desired voltage gain, it could be necessary to find the relationship between the normalized frequency fn and the duty ratio D for the given resonant parameters and loads. The proposed converter is able to control the output voltage by changing the duty ratio D. In the steady-state operation, Cr, Lr and Lm resonate after switch turn-off, so the turn-off time should be maintained until the voltage across Cr becomes the input voltage Vin at the end of mode 4. At that time, the switch voltage vSW is zero. Therefore, when the switch turn-on time for regulating the output voltage Vo is determined, the switching frequency fSW is determined dependently.

Fig. 5 shows the switching frequency and the switch turn-on time change for load variations with arbitrary resonant parameters. With light load, the switch turn-on time is reduced, so the switch period is reduced. On the other hand, the heavier the load, the larger the switch turn-on time. Consequently, the switch period is larger. Table 1 shows the variation trend of the switching frequency and turn-on time while the load changes.

Fig. 5.Variation of fSW and tON according to loads

Table 1.Variation trend of the switching frequency and turn-on time

Fig. 6 shows the simulation results on the relationship between the duty ratio D and the normalized frequency fn while the load varies from 25% to 133% under all given circuit conditions as shown in Fig. 6. The higher duty ratio implies the heavy load and the lower duty ratio does the light load. It could be found that an identical pattern between the duty ratio D and the normalized frequency fn exists regardless of circuit conditions. By using the least squares method (LSM), the normalized frequency fn has been approximated as a function of the duty ratio D:

Fig.6.Correlation between fn and D for different circuit parameters

The switch of the proposed converter is stressed by the voltage charged in the resonant capacitor Cr through the resonance between Cr, Lr, and Lm during switch turn-off. The resonant capacitor voltage vCr can be solved by calculating the maximum resonant current ICrmax. At the end of mode 4, the resonant capacitor voltage vCr reaches the input voltage Vin when the maximum resonant current decreases to iLr(t0).

By the law of energy conservation, the maximum current of the resonant capacitor Cr during mode 4 is given by:

The maximum capacitor voltage VCrmax normalized by the input voltage Vin is derived from (24):

Fig. 7 shows the maximum capacitor voltage according to load variations under some circuit conditions. As mentioned earlier, higher load increases the duty ratio D, leading to reduction of the normalized frequency fn. Therefore, the larger the load is, the higher the maximum value of the resonant capacitor voltage is. As shown in Fig. 7, the value of the resonant capacitance Cr and/or the value of the magnetizing inductance Lm to reduce the inductance ratio λ should be large.

Fig. 7.VCrmax normalized by Vin according to load variations

2.3 Control strategy

As analyzed in the previous section, the switching frequency of the proposed converter is dependent on the duty ratio. Therefore, the proposed converter requires a PWM controller that can generate a driving pulse with the switching frequency corresponding to the duty ratio variation in order to regulate the output voltage and satisfy ZVZCS operation for loads.

Fig. 8 shows the electric diagram with the proposed control strategy. In Fig. 8(a), the control block can be divided into three parts. First, the A-Part generates the zero current detecting signal (A signal), which evaluates the zero crossing of the switch current sensed from the shunt resistor. This A signal is used for the switch to be activated in the A-Region shown in Fig. 8(b) for ZVS operation. Second, the B-Part is the voltage controller which determines the conduction time value to keep the switch current flowing for the required output voltage. A type II compensator is used to amplify the difference between the feedback value and reference value. The determined time value is entered into the controlled monostable device. The monostable device is triggered at the falling edge of the A signal and then generates the pulse signal (B signal) corresponding to the conduction time value. Finally, the OR gate outputs the final PWM signal to drive the switch by combining both the A signal and the B signal. By using the proposed control strategy, the switching frequency and turn-on time can be appropriately controlled to regulate the output voltage.

Fig. 8.Electric diagram with the proposed control strategy: (a) Proposed circuit with control block and (b) PWM strategy

3. Experimental Results

3.1 Operation examination

In order to verify the theoretical analysis of the proposed converter, the hardware circuit shown in Fig. 1 is implemented. The input voltage, output voltage, and the maximum power are specified as Vin = 310 V, Vo = 12 V, and Pomax =72 W, respectively. The main components and parameters of the prototype used for experiments are presented in Table 2.

Table 2.Components and parameters for experiment

As shown in section III, the switch is stressed by the voltage charged in the resonant capacitor Cr through the resonance between Cr, Lr, and Lm during switch turn-off. The circulating energy should be minimized to reduce the switch voltage stress. An 900-V n-channel MOSFET has been adopted as the switch of the proposed converter. In order to restrict the peak value of the switch voltage less than 900 V, the resonant and magnetizing inductances Lr and Lm are selected with a inductance ratio λ = 0.38. In addition, it is desired that the resonant capacitance Cr be more than 15 nF.

Fig. 9 shows the experimental waveforms of the main components such as the switch voltage vSW, the switch current iSW, the rectifying diode voltage vD, and the rectifying diode current iD at 100 % and 30 % loads. All of the active semiconductors of the proposed converter are turned on and turned off under soft-switching conditions. At 100 % load, the switching frequency fSW is about 69 kHz for the duty ratio D = 0.4. At that time, the maximum value of the switch voltage is 834 V. It coincides with the theoretical results VSWmax = 836 V obtained from (25) with parameters in Table 2. Fig. 10 shows the efficiency comparison between the conventional flyback converter and the proposed converter. Though the proposed converter is voltage-stressed by the quasi-resonance compared to the flyback converter, the proposed converter achieves soft switching when all semiconductors such as the switch and rectifying diode are turned on and turned off. This implies that the switching losses are reduced, leading to increased system efficiency. As shown in Fig. 10, the efficiency of the proposed converter using an MOSFET as a switch is higher by up to 2 % compared to the conventional converter under light load. It can be shown that the overall efficiency is higher than the flyback converter. However, when an IGBT is adopted as a switch of the proposed converter, the efficiency is reduced. This results from the conduction loss increased by a high saturation voltage when the switch is on. Roughly, it could be estimated that the conduction loss of the IGBT is about 3 times higher than that of the MOSFET from the datasheets. Furthermore, the tail current of the IGBT at turn-off increases the switching loss, leading to deterioration of the system efficiency. It is shown that the MOSFET is the most suitable switching device.

Fig. 9.Main component waveforms: (a) 100 % load and (b) 30 % load: vSW (300 V/div), iSW (3 A/div), vD (20 V/div), and iD (10 A/div).

Fig. 10.Efficiency comparison according to switch devices

3.2 Electromagnetic interference examination

Electromagnetic emission is a critical concern in designing an SMPS for TVs. The conducted emission of the proposed prototype system is compared to the conventional flyback system applied to TVs.

The EMI filter specifications of both systems are identical. A test has been carried out to meet the CISPR Pub. 22 Class B Conducted Emissions Limit. The test results for the proposed converter and the flyback converter at the maximum output power Pomax=72 W are given in Fig. 11. The emission electrical fields of both systems below 10 MHz are confirmed to be lower than the standard reference level. However, at the frequency range of 1-30 MHz, the electrical field of the proposed converter system is much lower than that of the conventional flyback converter system. Even though it is known well that the conducted emissions higher than 10 MHz can be relatively solved by slowing down the turn-off speed of the main MOSFET. But for frequency around 1 MHz, the proposed converter shows outstanding advantages. The measured quasi-peak levels of the flyback converter system were 58.9 dBμV at 14.23 MHz and 61.2 dBμV at 25.6 MHz. On the other hand, the measured levels of the proposed system were 57.6 dBμV at 11.74 MHz and 50.1 dBμV at 27.1 MHz. From the results, it can be estimated that the radiated emission performance of the proposed converter will be superior to the conventional flyback converter by the lower di/dt and dv/dt.

Fig. 11.CE test results: (a) Flyback converter and (b) Proposed converter.

4. Conclusion

A single-switch ZVZCS quasi-resonant CLL isolated converter for low-power and low-priced electric appliances has been proposed. The proposed converter achieves the soft switching of all semiconductors during switching transition. Though the circulating current increases during the switch turn-off, the proposed converter provides the following advantages:

1) Improved efficiency by reducing switching losses through the soft switching of all semiconductors; 2) Low EMI above 1 MHz complying with the CISPR 22 Class B Conducted Emissions limit.

The proposed converter was validated with experimental results by implementing a 72-W prototype converter. The proposed converter shows high efficiencies for loads. The efficiency under light load is higher by more than 2 % compared to the conventional flyback converter.